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Project Title:

Switch Mode Power Supplies:
 






 Description:

Introduction - Some Definitions.

Switch Mode Power supplies are the present state of the art in high efficiency power supplies. typical series-regulated linear power supplies maintain a constant voltage by varying their resistance to deal with input voltage changes or load current demand changes. The linear regulator will, therefore, tend to be very inefficient. The switch mode power supply, however, uses a high frequency switch (in practice a transistor) with variable duty cycle to take care of the output voltage. The output voltage variations caused by the switching are filtered out by an LC filter.

SMPSs can be used to decrease a supply voltage, even as linear supplies do. not like a linear regulator, however, an SMPS can also give a step-up function and an inverted output function. Typical applications are given below.

Typical application for a ste-down switching regulator:

Generation of 5V for TTL-based circuits from a 12V battery (particularly suitable if the 12V battery has limited capacity, as switching regulators are far more efficient than linear regulators).

Typical application for a step-up switching regulator:

Generation of 25V from a 5V supply in an EPROM programmer.

Typical application for an inverting switching regulator:

Generation of a double-ended supply from a single-ended for OP-AMP.

Generation of a negative bias for MOS devices eg Dynamic RAMS.

The term switch mode regulator is used to describe a circuit which takes a DC input and provides a DC output of the same or opposite polarity, and of a lower or higher voltage. Switch Mode regulators use an inductor and there is no input to output regulation.

The term switch mode converter is used to describe a circuit which takes a DC input and provides a single or multiple DC outputs, again of same or opposite polarity and lower or higher voltage. Converters use a transformer and may provide input to output isolation.

The term Switch Mode Power Supply or SMPS is used to describe switch mode regulators and converters.


Forward and Flyback Circuits.

When discussing SMPS circuits, the different topologies are often referred to as 'Forward' or 'Flyback' A Feed Forward SMPS circuit will supply energy to the output capacitor when the switching element (transistor) is switched on. A Flyback SMPS circuit transfers energy (from an inductor) to the output capacitor when the switching element (transistor) is switched off.


The Buck Regulator (or Forward Regulator).




The Current-Boosted Buck Regulator.



The current-boosted buck convertor uses a transformer to extend output current higher than the maximum current rating of the switch (which is a transistor in a practical circuit). The current-boosted circuit will thus at the expense of increased  switch voltage throughout switch-off time. the rise in maximum output current over a typical buck convertor is equal to input voltage divided by output voltage, and turns ratio times the input-output differential. as an instance, during a 15V to 5V current-boosted buck convertor, with a 1:4 turns ratio, the rise in output current is double: 15/(5+1/4x15-5), or 2. this is often a 100 increase in output current. However, the maximum switch voltage for a current-boost buck is increased  from input voltage to input voltage and output voltage divided by the turns ratio. using the 15V to 5V convertor, the maximum switch voltage is 15+5/turns ratio, or 15+5/0.25=35.

The Boost Regulator.



In a forward (or buck) regulator power is endlessly provided to the outlet\filter capacitor. in a boost regulator, however, energy is pumped up in a very cyclic manner. The filter capacitor thus needs to be of a higher value. The boost regulator, just like the flyback regulator, pumps energy into the outlet filter capacitor in a very cyclic manner, and it's thus desirable to work within the discontinuous mode with a fixed peak current through the inductor.

The diode conduction time in a boost regulator, in contrast to the flyback regulator, isn't fastened, but varies with the input voltage.
Ipk = 2 x Iout,max x (Vout / Vin,min) Td x Ipk) / (Vout - Vin)

Output voltage is regulated by controlling the duty cycle.
Vout = ((Ton / Tdon) + 1) x Vin

Ripple voltage is directly proportional to diode conduction time.
Tdon max = (L x Ipk) / (Vout - Vin,max)

The Current-Boosted Boost Regulator.



The Flyback Regulator

The flyback regulator circuit shown below can be used as a step-up or step-down circuit.



In a forward (or buck) regulator power is continuously provided to the outlet filter capacitor. in a flyback regulator, however, energy is pumped-up in a very cyclic manner. The filter capacitor thus needs to be of a better value.

Flyback regulators will operate in one of two modes: Continuous or Discontinuous. In Continuous mode, a large value of inductance is employed such that the current within the inductance never falls to zero. In Discontinous mode, the current within the inductance falls to zero before the switch closes. typically the circuit is intended such that at worst case conditions (max output current, min input voltage) this only falls to zero for a rapid, i.e. as shortly because the diode stops conducting the switch is closed.

For a flyback regulator, the peak current is given by:

Ipk = (Vin x Ton)/L
where:
Ipk = Peak current - Amps
Vin = Input voltage - Volts T conduction time - Seconds
L = Inductance - Henries

The conduction time of the diode (which might or might not be identical because the off time of the switch) is given by:
 Td x L) / Vout
where:
Ipk = Peak current - Amps
Vout = Output voltage - Volts Td conduction time - Seconds
L = Inductance - Henries

The output power from a flyback regulator is given by:

Pout = Vout x Iout = 0.5 x L x Ipk^2 x f
where:
Iout = Average output current - Amps
Vout = Average output voltage - Volts Td conduction time - Seconds
L = Inductance - Henries
Ipk = Peak current - Amps
f = Frequency of operation - Hertz

Iout is additionally the average current through the diode (since all output current must flow through D:

Iout = (Ipk / 2) x (Tdon x f)
where:
Iout = Average output current - Amps
Ipk = Peak current - Amps Td conduction time - Seconds
f = Frequency of operation - Hertz

The reverse polarity output voltage is given by:

Vout = (Pout x Rl)^0.5

or:

Vout = Ipk x ((L x f x Rl)/2)^0.5 where:
Vout = Average output voltage - Volts
Pout = Average Output power - Watts
Rl = Load Resistance - Ohms
Ipk = Peak current - Amps
L = Inductance - Henries
f = Frequency of operation - Hertz

The output voltage of the circuit will be regulated by operating the circuit at a set frequency and ranging the electronic transistor duty cycle. However, due to the pumping action, the output voltage sags whereas the transistor switch is on and rises once the transistor is off. This makes the circuit tough to manage in a very fixed frequency manner. a much better approach to controlling the flyback device once in operation in the discontinuous mode is to possess a hard and fast peak current within the inductance and therefore fixed anode conduction time. The transistor switch 'on' time will then be varied reciprocally to any changes within the output voltage. this provides rise to the circuit having variable frequency of operation.


The Cúk; SMPS.

In 1978, the U.S.A. Patent Office granted Dr Slobodan Cúuk; of CalTech (pronounced Chook) a patent for the design of a brand new SMPS topology. the advantages of his new topology embrace enlarged efficiency, low input and output current ripple, least RFI and little size and weight. The series of diagrams below shows how the Cúuk; circuit is derived.

Firstly, consider a boost converter followed by a buck converter:



If the diodes are thought of as switches, then the two switch-and diode sets could be replaced with a DPDT switch:



The DPDT switch and shunt capacitor can be replaced with an SPST switch and series capacitor, providing that a reverse of output polarity is accepted:



The circuit is an example of a Cúuk; SMPS. A practical realisation could be as follows:



The DC voltage transformation ratio M is M=D/D', where D is the duty ratio (fractional on-time) of the electronic transistor switch operated at a switching frequency 1/Ts, and D'=1-D is the complementary duty ratio (fractional offtime). For a DC input voltage Vi, the output voltage is Vo=MVi. The convertor therefore encompasses a step down ratio for D<0.5 and a step-up ratio for D>0.5. the other principal feature is that each the input and output currents are non-pulsating, each being smoothed by the input and output inductors. These inductors additionally eliminate current surges within the electronic transistor switch at power-on and power-off, that is commonly a troublesome drawback to solve.

Theoretically, the output capacitance C2 isn't required, however is sometimes enclosed to soak up the load current fluctuations. The Cúuk; convertor is exclusive as a result of energy transfers capacitively from the input to the output, rather than inductively as in all previous converters.

A capacitor of 1uF charged to 50V contains a stored energy of 1.25mJ, equal to an inductor of 2.5mH passing 1A. the size of a 1uF 50V capacitor, however, is significantly smaller than a 2.5mH 1A choke. so capacitive energy transfer is more practical on a per unit size or weight basis than inductive energy transfer.

Because the input and output inductance currents are basically constant, the switching current is confined entirely within the convertor within the transistor-coupling capacitor-diode loop. With careful layout, this loop can be created physically tiny, which can cut back the radiated RFI from the field. additionally, the voltage and current waveforms during this convertor are significantly clean, with little ringing or overshoot; little snubbing is needed.

Cúuk; References

    S. Cúuk; and R.D. Middlebrook, "A New Optimum Topology switch DC-to-DC Converter", IEEE Power electronics Specialist Conference 1977 pp 160-179.
    S. Cúuk; and R.D. Middlebrook, "Coupled inductor and other Extensions of a new Optimum Topology switching Converter", IEEE Industrial Applications Society Annual Meeting, 1977 pp 1110-1126.
    S. Cúuk; and W.M. Polivka, Analysis of Integrated magnetics to Eliminate Current Ripple in switching Converters, Power Conversion International Proceedings, 1983 pp 361-386.




The Forward Converter.



The 'extra' winding of a forward converter's transformer ensures that at the beginning of a switch conduction, the net magnetization of the transformer core is zero. If there have been no additional winding, then after some cycles the transformer core would magnetically saturate, inflicting the primary current to rise excessively, therefore destroying the switch (ie transistor).

The diode on the secondary that's connected between the 0V line and also the junction of the inductance and rectifying diode is usually known as the 'flywheel diode'.

Waveforms for the forward converter are shown below:



The output voltage of a forward converter is equal to the average of the waveform applied to the LC filter and is given by:

Vout = Vin x (n2/n1) x (Ton x f)
where:
n2 = secondary turns on T1
n1 = primary turns on T1
Ton = conduction time of switch
f = frequency of operation


The Flyback Converter.



The output voltage for a flyback device (trapezoidal current flow operation) is also calculated as follows:

Vout=Vin x (n2/n1) x (Ton x f) x (1/(1-(Ton x f)))
where:
n2 = secondary activates T1
n1 = primary activates T1 T time of Q1

The control circuit monitors Vout and controls the duty cycle of the drive waveform to Q1.

If Vin will increase, the control circuit can cut back the duty cycle consequently, therefore as to maintain a constant output. Likewise if the load is reduced and Vout rises, the control circuit can act within the same manner. Conversely a decrease in Vin or increase in load, will cause the duty cycle to be raised.

It will be seen that the output voltage changes once the duty cycle, Ton x f, is changed. but the relationship between the output voltage and duty cycle isn't linear, as was the case with the forward converter, however instead it's a hyperbolic function.

The current flow in a flybcak converter will have either trapezoidal or sawtooth characteristics, as seen below. The trapezoidal current characteristic is thanks to the shift electronic transistor turning on again before the secondary current has dropped to zero. while the sawtooth characteristic is thanks to the secondary current falling to zero and there being a period of 'dead time' once there's no current flow in either secondary or primary.




The Push Pull Converter.



The push pull convertor belongs to the feed forward convertor family. With reference to the diagram above, when Q1 switches on, current flows through the 'upper' 1/2 T1's primary and also the magnetic field in T1 expands. The increasing magnetic field in T1 induces a voltage across T1 secondary, the polarity is such that D2 is forward biased and D1 reverse biased. D2 conducts and charges the output capacitor C2 via L1. L1 and C2 form an LC filter network. once Q1 turns off, the magnetic field in T1 collapses, and when a amount of dead time (dependent on the duty cycle of the PWM drive signal), Q2 conducts, current flows through the 'lower' 1/2 T1's primary and also the magnetic field in T1 expands. now the direction of the magnetic flux is opposite to that created once Q1 conducted. The increasing magnetic field induces a voltage across T1 secondary, the polarity is such that D1 is forward biased and D2 reverse biased. D1 conducts and charges the output capacitor C2 via L1. after a amount of dead time, Q1 conducts and also the cycle repeats.

There are two important considerations with the push pull converter:

1. Both transistors must not conduct together, as this would effectively short circuit the supply. Which means that the conduction time of each transistor must not exceed half of the total period for one complete cycle, otherwise conduction will overlap.
2. The magnetic behaviour of the circuit must be uniform, otherwise the transformer may saturate, and this would cause destruction of Q1 and Q2. This requires that the individual conduction times of Q1 and Q2 be exactly equal and the two halves of the centre-tapped transformer primary be magnetically identical.

These criteria must be satisfied by the control and drive circuit and the transformer.

The output voltage Vout equals the average of the waveform applied to the LC filter:

Vout = Vin x (n2/n1) x f x (Ton,q1 + Ton,q2)
where:
Vout=Average output voltage - Volts
Vin=Supply Voltage - Volts
n2=half of total number of secondary turns
n1=half of total number of primary turns
f = frequency of operation - Hertz
Ton,q1 = time period of Q1 conduction - Seconds
Ton,q2 = time period of Q2 conduction - Seconds


The control circuit monitors Vout and controls the duty cycle of the drive waveforms to Q1 and Q2.

If Vin will increase, the control circuit will cut back the duty cycle accordingly, therefore as to maintain a constant output. Likewise if the load is reduced and Vout rises the control circuit will act within the same method. Conversely, a decrease in Vin or increase in load, will cause the duty cycle to be increased. The diagram below shows associated waveforms from the push pull convertor.



The Half Bridge Converter



The half bridge convertor is comparable to the push pull convertor, however a centre tapped primary isn't needed. The reversal of the field is achieved by reversing the direction of the first winding current flow. this kind of convertor is found in high power applications.

For the half bridge convertor, the output voltage Vout equals the average of the waveform applied to the LC filter

Vout = (Vin/2) x (n2/n1) x f x (Ton,q1 + Ton,q2)
where
Vout=Output Voltage - Volts
Vin=Input Voltage - Volts
n2=0.5 x secondary turns
n1=primary turns
f = operating frequency - Hertz
Ton,q1 = Q1 conduction time - Seconds
Ton,q2 = Q2 conduction time - Seconds

Note that Ton,q1 = Ton,q2 which Q1 and Q2 are never conducting at identical time.

The control circuit of a half bridge convertor is comparable to that of a push-pull convertor.


The Full Bridge Converter



The full bridge converter is comparable to the push pull device, however a centre tapped  primary isn't needed. The reversal of the magnetic field is achieved by reversing the direction of the first winding current flow. this kind of converter is found in high power applications.

For the full bridge converter, the output voltage Vout equals the average of the waveform applied to the LC filter

Vout = Vin x (n2/n1) x f x (Ton,q1 + Ton,q2)
where
Vout=Output Voltage - Volts
Vin=Input Voltage - Volts
n2=0.5 x secondary turns
n1=primary turns
f = operating frequency - Hertz
Ton,q1 = Q1 conduction time - Seconds
Ton,q2 = Q2 conduction time - Seconds

Diagonal pairs of transistors will alternately conduct, therefore achieving current reversal within the transformer primary. this can be illustrated as follows - with Q1 and q4 conducting, current flow are going to be 'downwards' through the transformer primary, and with Q2 and Q3 conducting, current flow are going to be 'upwards' through the transformer primary.

The control circuit monitors Vout and controls the duty cycle of the drive waveform to Q1, Q2, Q3 and Q4.

The control circuit operates within the same manner as for the push-pull converter and half-bridge converter, except that four transistors are being driven instead of 2.


Control.



Driving the power stages and taking care of regulation is that the control circuit shown above, that shows a simplified circuit as an example the basics of operation. The control circuit itself needs power to work, and this is provided by the block selected 'Aux. Supply'. The auxiliary supply could take many forms, and in this instance it derives its power from the 240V AC mains supply input {to provide|to offer|to produce} the low voltage auxiliary supply. The circuitry needed to perform the control functions will usually be within the variety of an integrated circuit. In figure 15b the blocks tagged with a letter ('a' to 'm') are contained in an integrated circuit, while the other discrete components are external to the device. Block 'a' is a extremely stable oscillator that produces a sawtooth wave, timing components R1 and C1 set the operative frequency. A stable reference voltage (VREF) is provided by block 'b', this is utilized by the error amplifier and other circuitry. control of the output pulse width is under control of the Dead Time Comparator, 'c', and therefore the Pulse width Modulator (PWM) Comparator, 'e'. These 2 comparators each compare a control voltage with the sawtooth oscillator wave. The output of the two comparators is ORed along, 'f', before being fed to the output control circuitry. The Dead Time control Comparator is employed {to provide|to offer|to produce} a 'soft start'; this brings the output of the power supply up to its operating level over a brief amount of time (50ms) once activate. This reduces stress on the components within the power stages at switch-on. The time constant for the 'soft start' is provided by R2 and C2. The Error amplifier, 'd', is accountable for comparing the output voltage of the power supply with a reference. If the output voltage tries to deviate from its correct value, because of changes in either load and/or line voltage, the error amplifier output changes to take care of the right output voltage. The voltage at the output of the power supply is monitored at the point marked 'FB' on Figure 15a, this voltage is coupled to the error amplifier circuitry via an Opto Isolator (Figure 15b). The Opto Isolator is needed to take care of isolation between the low voltage output and therefore the live power stages. The opto isolation circuitry has been shown as a block for clarity. The output from the Opto Isolator is fed to the non-inverting input of the Error amplifier via a potential divider, R6 and R9, wherever it's compared to the reference voltage provided by another potential divider, R3, R4 and RV1. RV1 serves to regulate the output voltage of the power supply to the desired level. C3 decouples the reference voltage. The gain of the error amplifier is set by the network of components, R5, R8, C4 and R7. The error amplifier is basically a non-inverting operational amplifier circuit with the low frequency gain being set by R5 and R8.

Error Amplifier LF Gain = 1 + (R8/R5)

The combination of inductive and capacitive parts within the power stages will cause phase shift, and at certain frequencies it's a possibility that the feedback loop might become unstable and break into oscillation, which is, to mention the smallest amount, extremely undesirable! the solution to the present is to supply frequency/phase compensation; this is provided by C4 and R7. because the frequency rises, the gain of the amplifier is reduced, therefore guaranteeing stability. The gain of the error amplifier at a given frequency is thus:

(where f is the frequency)

Impedance of frequency compensation network:

Xc = 1/(2 x pi x f x C4)

Z = (R7^2 + Xc^2)^0.5

Total Feedback Impedance:

Zf = (Z x R8) / (Z + R8)

Error amplifier Gain at Frequency f = 1 + (Zf/R5)

The integrated circuit utilized in this instance has 2 outputs, and may be 'programmed' to use these for either push-pull or single ended applications. choice of the specified mode is achieved by taking the inputs of the 2 AND gates (h) and (i) to either VREF for push-pull or 0V for single ended. once in push-pull mode, the frequency of the drive signal is half that of the oscillator, this is because of the action of bistable (g). The bistable ensures that the 2 output transistors (l) and (m) never conduct together which the duty cycle of every drive signal never exceeds 500th. the 500 maximum is a requirement in this sort of forward converter power supply. For this reason the push-pull mode is chosen, even though just one output transistor (l) is employed. NOR gates (j) and (k) steer the PWM drive signal to the output transistor under the control of the bistable outputs. R11 provides an emitter load for the output transistor.


Off-Line SMPS Circuits

A switch mode power supply that gives a low voltage isolated output from a mains source is often referred to as an 'Off-Line SMPS'. A typical block diagram of such a supply is shown below:



The filter shown on the left of the diagram is necessary to prevent the supply from causing interference on the mains wiring. It can also help to protect the SMPS circuitry from voltage spikes (or power surges) on the mains supply.

The filter shown on the left of the diagram is necessary to prevent the supply from causing interference on the mains wiring. It can also help to protect the SMPS circuitry from voltage spikes (or power surges) on the mains supply.

A typical power section of such a circuit is shown below:


The reservoir capacitor potential would be close to 340V for a 240V mains supply. this kind of supply is typically designed to be powered from either 120V or 240V mains. The SMPS is ready for the appropriate voltage with either a switch or a wire link. The switch activates a capacitive voltage doubling circuit once set for 120V. resistor R1 is a low price resistor of 2 to 4 ohms that protects the circuit from the current surge that may otherwise occur at switch-on as C1 is charged. Q1 is a high voltage power MOSFET that's used as a high speed switch to produce current pulses to the ferrite cored, high frequency transformer T1. The frequency of switching is usually within the range 25 to 250kHz. R2 and C2 form a snubber network to cut back voltage spikes and switching noise. Regulation is achieved by watching the output voltage at the point marked 'FB' and adjusting the pulse width of the Q1 gate drive. Fuse FS2 is for short-circuit \ overload protection. FS2 is typically replaced by a current sensing circuit which is able to shut the drive circuitry down within the event of excess current.


Switch Mode Converters.

In a mains-driven linear power supply, a mains frequency power transformer is used for isolation, and then a rectifier and linear regulator are used to provide the output voltage.

In a mains-driven SMPS, isolation, regulation and high-efficiency are combined into one. The SMPS uses a less bulky high frequency transformer typically operating at 25kHz to 250kHz (though low-power operation up to 1MHz is not unheard of).

The transformers and inductors used for SMPSs have ferrite dust cores as opposed to the laminated iron cores of their lower frequency counterparts. SMPS transformers generally have fewer turns in the windings than the mains frequency transformers.


SMPS RFI & EMI

Switch mode power supplies are infamous for making rfi and emi. Lowpass filters within the mains leads are important to cut back conducted interference. faraday screens between the transformer windings and around sensitive parts, together with correct field cancelling layouts on the circuit board, also are required to cut back e.m.i. and r.f.i. the issues of smoothing of sawtooth current waveforms puts a strain on capacitor style. The series inductance and resistance of normal electrolytic capacitors features a large effect on residual ripple and noise voltages at the outputs. For low powers and extremely low ripple and noise levels at the outputs, linear supplies can't be beaten. that's the main reason they hold their own there.


SMPS Integrated Circuits.

Mullard:
TDA2640
TDA2581
SGS
L4960

   Input Voltage Range: 9-50 VDC
    Output voltage Adjustable between 5 and 40 V
    Maximum output current: 2.5 A
    Maximum output power: 100 W
    Built-in soft start circuitry
    Stability of internal reference: +\- 4%
    Requires very few external components
    Duty Factor: 0-1
    High Efficiency: up to 90%
    Built-in thermal overload protection: chip shuts down when junction temp reaches 150 degrees C.
    Built-in current limiter for short-circuit protection

L4962 (16-pin DIL package. Output current up to 1.5A)
L4964 (special 15-pin enclosure. Output current up to 4A)

TL494
TL497
 The TL497 has an osillator with a fixed on-time, but variable frequency output. This gives minimal external component count. The switch-on time is controlled by the value of a capacitor connected between pin 3 and ground.
 



SMPS 'Hiccup' Mode

In switch-Mode Power Supplies the 'hiccup' mode is often used for limiting output current. If an overload occurs, the circuit turns off. After an interval it comes on - has a look, as it were; if the overload is still present, it immediately goes off again. In some designs, this happens a few times, and the supply then shuts down permanently until the overload is removed and the circuit reset.


SMPS Hold Up

Most offline switchers are designed to maintain a steady output over a few cycles of lost mains input. This can be achieved by sizing the input capacitor such that its voltage will not fall significantly during the power interruption. The time period over which the SMPS is capable of maintaining an output when mains power is lost is frequently known as 'hold up time'.









  














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